IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 33, NO. 6, NOVEMBER/DECEMBER 1997 1601
Switching Voltage Transient Protection Schemes
for High-Current IGBT Modules
Rahul S. Chokhawala, Member, IEEE, and Saed Sobhani
Abstract The emergence of high-current and faster switching
insulated gate bipolar transistor (IGBT) modules has made it
imperative for designers to look at ways of protecting these
devices against detrimental switching voltage transients that are a
common side effect of these efficient transistors. This paper will
discuss protection criteria for both normal switching operation
and short-circuit operation and will cover in detail some of the
protection schemes that were designed to address these problems.
Index Terms Insulated gate bipolar transistor (IGBT), protec-
tions circuits, resistor capacitor diode (RCD), short-circuit safe
operating area (SCSOA), short circuit, switching safe operating
area (SSOA), switching transients, voltage clamps.
I. INTRODUCTION
HEN a power device is abruptly turned off, trapped
energy in the circuit stray inductance is dissipated in
W
the switching device, causing a voltage overshoot across the
Fig. 1. Rated SOA curve and operating loci with and without switching
device. The magnitude of this transient voltage is proportional
voltage transient protection circuit.
to the amount of stray inductance and the rate of fall of turnoff
current. Large insulated gate bipolar transistor (IGBT) modules
switch high magnitude of currents in a short duration of time,
It is determined that the snubbers and clamps offer op-
giving rise to potentially destructive voltage transients. These
timized protection against voltage transients during normal
higher current modules normally consist of several IGBT chips
switching operation. Operation of a resistor capacitor diode
in parallel. Each individual chip switches its share of the load
(RCD) clamp circuit is described in detail. As illustrated in
current at a that is determined by the gate drive circuit.
Fig. 1, protection circuits allow faster yet safer operation by
The total current and seen by the external power circuit
containing operating loci within the boundaries of the rated
is the sum of currents and  s through each IGBT chip.
safe operating area (SOA).
The situation is at its worst when a short-circuit current is
Fault current shut-off transients are more effectively pro-
rapidly turned off to protect the IGBT. The  s produced
tected by considerably slowing the rate of fall of fault current.
could easily be a few thousand A/ s. If proper attention is not
Two novel protection schemes are introduced which protect
paid to minimize resulting switching voltage transients, any
IGBT s from potentially destructive voltage transients by
attempt to save IGBT s, by shutting them down under fault
slowing the rate of fall of fault current, only under fault
conditions, may destroy the device.
conditions. Circuit operations are analyzed and the test results
This paper discusses various protection schemes. A tran-
are illustrated. Usefulness of an active clamp is also discussed
sient voltage protection scheme optimized to protect IGBT s
in this section.
during normal switching operation may not protect the IGBT s
under fault current shut-off process. Separate schemes would
II. VOLTAGE TRANSIENTS DURING
normally be required to achieve both goals.
NORMAL SWITCHING OPERATION
As mentioned earlier, the magnitude of transient voltage
depends on the trapped energy in the circuit stray inductance,
Paper IPCSD 97 32, presented at the 1994 IEEE Applied Power Electronics
Conference and Exposition, Orlando, FL, February 13 17, and approved for also called  dc loop inductance As a preventive measure,
publication in the IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS by the
steps should be taken to improve the circuit layout. Usage of
Power Electronics Devices and Components Committee of the IEEE Industry
copper plates separated by a thin sheet of insulating material,
Applications Society. Manuscript released for publication May 9, 1997.
R. S. Chokhawala is with ABB Semiconductors AG, CH-5600 Lenzburg, tightening the  dc loop, and choosing source capacitance
Switzerland.
with inherently low self inductance are ways to lower stray
S. Sobhani is with International Rectifier Corporation, El Segundo, CA
inductances [1]. Decoupling capacitors, connected across the
90245 USA.
Publisher Item Identifier S 0093-9994(97)08440-5. module terminals, can also be used to achieve this goal. High-
0093 9994/97$10.00 © 1997 IEEE
1602 IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 33, NO. 6, NOVEMBER/DECEMBER 1997
Fig. 2. RCD voltage clamp.
Fig. 4. Turnoff waveforms with and without RCD snubber. Tested at: 400 V,
100 A, 25 C; Ls = 100 nH; VG=RG(o ) = -8 V/33 ; Csn = 0.22 F;
Rsn = 12 ; VCE: 100 V/div; IC=Isn: 20 A/div; Eo : 2 mJ/div.
also for reducing IGBT turnoff losses. During IGBT turn-on,
the snubber capacitor is fully discharged, and during turnoff,
it is charged. This circuit, unlike the circuit in Fig. 2 which
essentially acts as a clamp, reduces the rate of rise of voltage
across the IGBT at turnoff, imposing a softer switching and,
therefore, reducing losses in the IGBT. The losses in the
snubber, however, are substantially increased and are equal to
where is the voltage across snubber capacitor
at the end of turnoff process and is equal to the dc bus plus
Fig. 3. RCD charge/discharge snubber.
an allowable overshoot voltage.
Due to the dual purpose that the circuit in Fig. 3 serves, the
frequency polypropylene capacitors designed for low internal
tradeoffs involved are complex. Since this paper concentrates
lead inductance are found to be effective. Care should be
only on the switching voltage transient protection, discussion
taken in the selection of the decoupling capacitor value to
will be focused on the circuit in Fig. 2. The effects of this
avoid oscillations in the dc loop which otherwise may result
snubber on turnoff and turn-on will be discussed separately in
in excessive heating in the high-frequency capacitors. For
the following.
modules rated up to 100 A or so, decoupling capacitors may
provide optimal protection against voltage transients during
A. Turnoff
normal switching.
The RCD clamp of Fig. 2 acts as a voltage clamp. During
One other way to prevent high-voltage transients from
IGBT conduction period, the snubber capacitors are charged
occurring is to slow down the switching process, by choosing a
to the bus voltage. As the IGBT is turned off, the voltage
greater value of gate resistor. While this is an attractive method
across it, , rises rapidly. The circuit  dc loop stray
for the fault current turnoff protection, it is not practical for
inductance may cause to rise above the bus voltage.
protection against voltage transients during normal switching
As this occurs, the snubber diode is forward biased, and
operation, as the efficiency of device operation is adversely
the snubber is activated. The energy trapped in the stray
affected.
inductance now is diverted to the snubber capacitor, which
In the following discussion, more efficient ways of protect-
absorbs this incremental energy without substantial rise in its
ing devices will be presented.
voltage. The waveforms shown in Fig. 4 clearly illustrate the
turnoff behavior with and without the RCD clamp. The voltage
III. RCD SNUBBER AND CLAMP CIRCUITS
overshoot has been substantially reduced from 210 to only 50
Figs. 2 and 3 are two principal examples of RCD snubbers V. Initially, a small stray inductance in the snubber circuit
for high-current IGBT applications. While both circuits are causes to peak slightly above
employed to reduce transient voltages across switching de- Fig. 5 displays the waveforms generated for two different
vices, the charge/discharge snubber circuit in Fig. 3 is targeted stray inductances (100 and 340 nH). As illustrated in the
CHOKHAWALA AND SOBHANI: SWITCHING VOLTAGE TRANSIENT PROTECTION SCHEMES 1603
Fig. 6. Turn-on waveforms for an IGBT with no RCD snubber protection.
Tested at: 400 V, 100 A, 25 C; Ls = 240 nH; VG=RG(on) = 15 V/5.1 ;
Fig. 5. Turnoff waveforms with RCD snubber for two different Ls values
VCE: 100 V/div; IC: 20 A/div; Vdiode: 100 V/div.
(100 nH, 340 nH). Tested at: 400 V, 100 A, 25 C; VG=RG(o ) = -8 V/33
; Csn = 0.22 F; Rsn = 12 ; VCE: 100 V/div; IC: 20 A/div; Eo :
2 mJ/div.
figure, the initial peak which is dependent on the
stray inductance within the snubber circuitry is the same for
the two cases. The final voltage peak for the higher
inductance does reach a higher value as expected, since there
is more trapped energy diverted to the same snubber
capacitor. This value, however, is well within the voltage
rating of the device and only marginally influences the losses
in the IGBT, since it occurs when the current has reached to a
smaller value. The magnitude can be calculated from the
formulas given in the following section.
B. Turn-On
Fig. 6 displays the turn-on waveforms for an unprotected
IGBT with a gate resistor of 5.1 . The rapid rise in
the IGBT current (1200 A/ s) combined with the circuit stray
Fig. 7. Turn-on waveforms for an IGBT with no RCD snubber protection.
inductance (300 nH) caused the freewheel diode (FWD) to go
Tested at: 400 V, 100 A, 25 C; Ls = 240 nH; VG=RG(on) = 15 V/33 ;
VCE: 100 V/div; IC: 20 A/div; Vdiode: 100 V/div; Eon: 1 mJ/div.
through severe reverse-recovery process. As seen in the figure,
the FWD recovery voltage ( 630 V) actually exceeded the
rated voltage of the module. discharge current partially provides for the reverse-
In order to bring this voltage down to a safe value, the turn- recovery charge of the FWD, thus, the total current seen by
on was lowered by using a higher The results are is modified. This has a favorable effect on the magnitude
shown in Fig. 7. The increase in however, had profound of the reverse-recovery voltage transient.
effect in increasing the switching losses, as expected [2], [3]. The waveforms shown in Fig. 9 illustrate the snubber oper-
The RCD clamp shown in Fig. 2 is also effective in reducing ation. Notice the complete elimination of the voltage transient
turn-on voltage transients. As the IGBT current rises, the and, also, reduction in the oscillations following turn-on.
voltage loss causes the voltage across the positive Another interesting fact is that this waveform was generated
and negative terminals of the module, to drop by the same with of 0.5 , which reduced the energy losses from 2.41
amount (i.e., to The snubber capacitors mJ in Fig. 7 to 1.25 mJ, a savings of almost 50%. Therefore,
that were fully charged to now find a discharge path this snubber not only clamps the turn-on voltage transient, but
through the forward-biased FWD (note that the FWD is on, also enables the user to choose a value of that produces
freewheeling the load current), the IGBT, and the snubber minimal turn-on losses.
resistors. Fig. 8 shows the equivalent circuit during turn-on. Fig. 10 shows the effect of changing the snubber resistor
The snubber diodes are reverse biased and, therefore, not on turn-on waveforms. Lower  s provide for
shown. The current paths are shown in the figure. This snubber better snubbing action.
1604 IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 33, NO. 6, NOVEMBER/DECEMBER 1997
Fig. 8. Equivalent circuit under turn-on conditions. Low-side IGBT is
Fig. 10. Effect of changing Rsn values. Tested at: 400 V, 100 A, 25 C;
switched on.
Ls = 240 nH; VG=RG(on) = 15 V/33 ; Csn = 0.22 F; Rsn = 12 ,
33 ; Vdiode: 100 V/div; Itotal: 20 A/div.
Losses in the snubber resistor are
(3)
The snubber diode should be of fast and soft recovery
type to avoid severe oscillations following at turnoff. The
resistor should be of noninductive type to avoid oscillations
at turn-on.
IV. VOLTAGE TRANSIENTS DURING
FAULT CURRENT TURNOFF
The short-circuit current generated during fault conditions
can be up to five to ten times the rated current. Shutting off
such high currents too quickly can produce extremely high
 s that are potentially detrimental to the IGBT s [4].
The RCD clamp circuit discussed in the preceding section is
not as practical when it comes to protecting transient voltages
Fig. 9. Turn-on waveforms with RCD snubber. Tested at: 400 V, 100 A, 25
generated during short-circuit conditions. As seen from the
C; Ls = 240 nH; VG=RG(on) = 15 V/0.5 ; Csn = 0.22 F; Rsn =
12 ; VCE=Vdiode: 100 V/div; IC =Isn=Itotal: 20 A/div; Eon: 0.5 mJ/div; above expressions, the required snubber capacitor value is
Vab: 100 V/div.
proportional to the square of the device current. This means
that the capacitor required will have to be 25 100 times
larger than in normal switching operation. High-capacity high-
The value for the snubber components can be approximated
voltage snubber capacitors are large and expensive (as for
from the expressions given below, based on circuit stray
snubber capacitors required for GTO thyristors), making the
inductance switching frequency maximum switching
RCD scheme unattractive for high-current IGBT modules.
current turn-on current rise time dc rail voltage ,
Also, voltage clamps connected externally to the modules
and allowable peak voltage (see the Appendix).
do not address the problem of internal inductive voltage
The snubber capacitor is
spike.
In the following, we will discuss more practical methods.
(1)
This involves slowing down the turnoff of the IGBT s under
fault conditions.
The snubber resistor is
The IGBT fault current rate can be reduced by slowing the
turnoff gate voltage signal. The simplest way of achieving
(2) this is to increase the gate resistor, but this is inefficient,
CHOKHAWALA AND SOBHANI: SWITCHING VOLTAGE TRANSIENT PROTECTION SCHEMES 1605
Fig. 12. Short-circuit waveforms with resistive protection scheme for  fault
under load condition. Tested at: 280 V, 25 C; Ls = 240 nH; VG=RG1 =
-8 V/33 ; VCE: 100 V/div; VGE = 5 V/div; Isc: 200 A/div.
Fig. 11. Circuit for the resistive method.
since the tradeoff is increased switching losses during normal
conduction.
In order to address the above problem, two novel circuits
are introduced that, through electronic gate control, effectively
decrease the rate of fall only when a fault current is
sensed, thereby avoiding any losses during normal switching
operation. The first of these circuits utilizes a resistive method,
and the other uses a capacitive method. In the resistive method,
a considerably higher value of gate resistor is switched in,
in series with the IGBT gate. In the capacitive method, a
considerably higher value of external capacitor is switched
in, in parallel with the IGBT gate input capacitance.
Fig. 13. Short-circuit waveforms with no protection scheme for  fault under
load condition. Tested at: 280 V, 25 C; Ls = 240 nH; VG=RG1 = -8
V/33 ; VCE: 100 V/div; VGE = 5 V/div; Isc : 200 A/div.
V. RESISTIVE METHOD
The circuit in Fig. 11 is composed of de-sat sense diode
and a p-channel MOSFET to switch in higher value of resistor
as the discharge is now forced to take place through
upon occurrence of a fault.
The fault current fall rate is decreased accordingly. Note that,
Initially, when the IGBT is in the off state, the P-MOSFET
if the IGBT is turned off while the MOSFET is still on,
is turned off. During normal turn-on, a step rise in voltage
the circuit will not be effective, since is bypassed. The
is applied to the IGBT gate through and the inherent
above consideration places an upper limit on the discharge
body diode of the P-MOSFET. As after normal turn-
time constant of the MOSFET (to, for example, 5 s).
on delay period, drops to its low on-state level, diode
Figs. 12 and 13 display short-circuit switching waveforms
is forward biased and input capacitance of the P-MOSFET
with and without the protection circuit. The initial current
starts to charge up. During normal conduction, therefore, the
through the IGBT is 40 A. Upon occurrence of the fault, the
P-MOSFET remains gated on.
current shoots up to 800 A initially, but settles down to 600 A
During normal turnoff operation, the gate drive output
once the Miller effect on the gate voltage is diminished (see
voltage is switched to its low state. The P-MOSFET gate
gate waveforms). The MOSFET discharge time constant was
capacitance begins to discharge. The values of and
adjusted to be 2.5 s Fault current
are adjusted such that the MOSFET is kept on, at least until
was turned off after 6 s. As seen from the aforementioned
the IGBT turnoff is completed (for example, 1 s). Therefore,
figures, the IGBT gate discharge rate was considerably slowed
the IGBT turnoff losses are not affected.
by the addition of (200 ), thereby reducing the voltage
overshoot from 270 down to 60 V.
A. Fault Under Load
The Miller effect can be filtered out by bypassing
When a fault occurs during normal conduction, diode with a diode. The IGBT gate voltage is now clamped to
goes into blocking mode and the P-MOSFET input capacitance the gate drive output voltage. Fig. 14 displays the resulting
starts to discharge through resistors and The MOSFET waveforms. Compare the results to Fig. 12 (same protection
is turned off as its gate voltage drops below the threshold circuit without the bypass diode). The initial surge of
value. Thereafter, the rate of fall is reduced significantly current is eliminated. The slight increase in the turnoff voltage
1606 IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 33, NO. 6, NOVEMBER/DECEMBER 1997
Fig. 16. Short-circuit waveforms with no protection scheme for  hard fault
condition. Tested at: 400 V, 25 C; Ls = 240 nH; VG=RG1 = -8 V/33 ;
Fig. 14. Short-circuit waveforms with resistive protection scheme for  fault
VCE: 100 V/div; VGE = 5 V/div; Isc: 200 A/div.
under load condition, with bypass diode across RG1 . Tested at: 280 V, 25
C; Ls = 240 nH; VG=RG1 = -8 V/0 ; VCE: 100 V/div; VGE = 5
V/div; Isc: 200 A/div.
Fig. 15. Short-circuit waveforms with resistive protection scheme for  hard
fault condition. Tested at: 400 V, 25 C; Ls = 240 nH; VG=RG1 = -8
V/33 ; VCE: 100 V/div; VGE = 5 V/div; Isc: 200 A/div.
Fig. 17. Circuit for the capacitive method.
overshoot, as the is now bypassed during the IGBT
turnoff, can be compensated for by readjusting value.
is adjusted such that it
is not turned on prior to the IGBT turning on (for example, 1
B. Hard Fault
s). Therefore, during the conduction period, the MOSFET is
in its off state. Zener diode is selected to offset large
This is the case where the device turns on directly into a
(on) voltages. False triggering of the MOSFET is, therefore,
fault. The MOSFET is never turned on in this case. Therefore,
prevented.
the discharge path for the is through Figs. 15 and 16
When the gate drive is switched to its off state to turn
show the waveforms with and without protection circuit. The
IGBT off, MOSFET remains turned off. The normal switching
voltage overshoot was brought down to only 10 from 190 V.
operation is, therefore, not affected by the existence of this
Note that this circuit increases the effective gate bias
protection circuit.
impedance during IGBT off time. This effect imposes an
upper limit on , the value of which is governed by the
IGBT characteristics and gate bias voltage.
A. Fault Under Load
Once a fault occurs, the sense diode becomes reverse biased
VI. CAPACITIVE METHOD
and the MOSFET gate input capacitance is charged by the
The circuit in Fig. 17 is composed of de-sat diode used gate drive power, through the voltage divider provided by
to sense a fault condition, and an -channel MOSFET to and When the MOSFET turns on, capacitor
switch in a higher value of capacitor, in parallel with the is switched in, in parallel with the IGBT input capacitance.
IGBT input capacitance upon occurrence of a fault. A drop is seen in the IGBT gate voltage, since some charge
During normal turn-on after the normal turn-on delay, is removed to charge the capacitor which was initially
drops to its low on-state level and diode is forward biased. charged to the off-bias voltage. This, in turn, lowers the value
The MOSFET gate charge time constant of the momentarily, reducing the energy losses during
CHOKHAWALA AND SOBHANI: SWITCHING VOLTAGE TRANSIENT PROTECTION SCHEMES 1607
Fig. 18. Short-circuit waveforms with capacitive protection scheme for  fault
under load condition. Tested at: 280 V, 25 C; Ls = 240 nH; VG=RG(o ) =
-8 V/33 ; VCE: 100 V/div; VGE = 5 V/div; Isc: 200 A/div.
the short-circuit period. The IGBT discharge time constant
has increased, since it now includes capacitor in parallel
with the IGBT input capacitance. The fault current turnoff
is, therefore, slowed, and the transient voltage is brought
Fig. 19. Short-circuit waveforms with and without capacitive protection
scheme for  hard fault condition. Tested at: 340 V, 25 C; Ls = 240
down substantially. Fig. 18 shows the waveforms for the IGBT
nH; VG=RG(o ) = -8 V/33 ; VCE: 100 V/div; VGE = 10 V/div; Isc:
with the protection circuit. Compare this to the waveforms
200 A/div.
in Fig. 13 without the protection circuit. Once again, turnoff
voltage overshoot was reduced from 270 to 60 V.
B. Hard Fault
The operation of this protection circuit under hard fault
is the same as described above. The upper limit of the
MOSFET s gate charge time constant should be adjusted such
that the MOSFET is fully turned on before the fault current
is turned off (e.g., in less than 5 s). Fig. 19 displays the
waveforms with and without the protection circuitry. As seen
from the figure, the voltage overshoot was brought down from
160 to 50 V.
The functional usefulness of the circuit in Fig. 17 can
be increased by simple addition of Zener diode across
capacitor (indicated in the figure). The circuit now serves
the dual purpose of preventing turnoff voltage transient, as well
Fig. 20. Circuit for Zener clamp protection scheme.
as limiting fault current amplitude. The complete discussion
on such a fault current limiting circuit is presented in [5].
voltage across 600-V rated IGBT s was reduced from 580 V
VII. ACTIVE VOLTAGE CLAMP
to a safer level of 460 V.
While the previously discussed (resistive and capacitive)
The circuit in Fig. 20 contains avalanche diode in series
circuits are activated immediately upon commencement of a
with blocking diode connected between IGBT collector
fault, the circuit presently under consideration reacts only at
and gate. The avalanche diode is selected such that its voltage
turnoff. A slightest delay in regating of the IGBT would result
rating is less than the maximum allowable voltage at the
in a potentially dangerous voltage spike. It was found that the
IGBT module terminals. If this voltage limit is exceeded at
combined junction capacitances of and if high enough,
the turnoff, the avalanche current in would raise
quickens charging of the IGBT gate by providing a
the gate-emitter voltage above its threshold level, therefore
maintaining IGBT in conducting state. This feedback mecha- feedback. At the same time, these capacitances, if too high,
nism clamps to a safe value. The rate of decay of is would noticeably contribute to the adverse Miller effect under
then equal to where is the clamp voltage normal operating conditions. Proper selection of these com-
and is the dc-bus voltage. Fig. 21 illustrates operation of ponents is, therefore, extremely crucial to assure successful
this clamp circuit. For comparison, results obtained without operation of this circuit. Also note that the value of
this protection circuit are shown in Fig. 22. The peak turnoff if too low, will make operation of this circuit less effective.
1608 IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 33, NO. 6, NOVEMBER/DECEMBER 1997
stray inductance. For of 550 V and of 400 V, the
additional losses per pulse in the IGBT are almost four times
that value.
The additional losses inflicted by the slower turnoff is not
a major consideration during the fault operation, as it is only
a one-time operation. Usefulness of this circuit is, therefore,
restricted to fault current protection.
VIII. CONCLUSIONS
The problem of switching voltage transients is an important
subject that cannot be ignored, especially in applications where
high-current IGBT modules are used. This paper has discussed
principal sources of voltage transients. Protection schemes
that were designed and tested on high-current IGBT modules
under normal switching operation and fault condition were
described.
An RCD clamp circuit was the focus for the overvoltage
protection during normal switching operation. This low-loss
circuit offers effective protection against voltage transients
during normal turn-on and turnoff switching.
Fig. 21. Short-circuit waveforms with Zener clamp protection scheme
for  hard fault condition. Tested at: 300 V, 25 C; Ls = 240 nH;
Voltage transients during fault current shutoff are more
VG=RG(o ) = -5 V/33 ; VCE: 100 V/div; VGE = 5 V/div; Isc: 200
effectively protected by slowing the rate of fall of fault current.
A/div.
Two novel protection schemes, resistive and capacitive tech-
niques, were introduced. These circuits, through electronic gate
control, slow the rate of decay of gate voltage, thereby slowing
the rate of fall of fault current. Operational characteristics of
an active clamp circuit were also discussed in this section.
Circuit operations were analyzed, and the test results were
illustrated.
APPENDIX
The expressions (1) (3) are derived as follows.
At turnoff (see Fig. 2), as one of the two conducting IGBT s
is gated off, collector to emitter voltage rises to the dc-
bus voltage Beyond this point, load current freewheels
through the diode across the other IGBT. The stray inductances
, however, prolong flow of current in the  dc loop. Two
components of currents make up for the current in
They are IGBT turnoff current and the snubber current
as shown in Fig. 4.
For simplicity of calculations, it is assumed that the IGBT
turns off  instantly, i.e., is equal to This assump-
tion is justified on the grounds that it only renders a somewhat
Fig. 22. Short-circuit waveforms without Zener clamp protection scheme
conservative estimate of snubber capacitor value.
for  hard fault condition. Tested at: 300 V, 25 C; Ls = 240 nH;
VG=RG(o ) = -5 V/33 ; VCE: 100 V/div; VGE = 5 V/div; Isc : 200
The equations governing various circuit variables are as
A/div.
follows:
Except for unclamped inductive load applications, this cir-
i.e.,
cuit is not appropriate to protect against voltage transients
during normal switching operation, as substantial losses are
(a)
incurred in the IGBT due to operation of the voltage clamp.
The losses per pulse are given by the following equation:
i.e.,
The value is the energy trapped in the  dc loop (b)
CHOKHAWALA AND SOBHANI: SWITCHING VOLTAGE TRANSIENT PROTECTION SCHEMES 1609
From (a) and (b), The snubber discharge current is
(g)
(c)
where
The following is the solution to the above equation:
(h)
(d)
where
where is the load current at the turnoff. The current is,
thus, a cosine function. In Fig. 4, it can be observed that
(i)
the combination of and does follow cosine wave
shape.
and is the rise time, specified under inductive load condi-
Differentiating (d),
tions. From (g) (i),
(e)
(j)
From (b) and (e),
The losses in at turn-on are
(f)
(k)
is at maximum when Therefore,
where the snubber discharge time, is approximated to be
the interval between the beginning of the current rise and the
maximum desired voltage across IGBT
point where current reaches its peak value :
(g)
(l)
From (g), snubber capacitor
Combining (k) and (l),
(1)
(m)
The snubber capacitor is, therefore, charged to at the end
From (f) and (m), the total losses in the snubber resistor are
of turnoff. Before the next turnoff event, i.e., later,
should discharge back to its initial value of The snubber
resistor selected according to the following expression
(3)
fulfills the above requirement:
(2)
REFERENCES
[1] H. Vetter,  High performance capacitors for low-inductance circuits,
The losses in at turnoff are, therefore, given by
Power Conversion, June 1991.
[2] C. Aniceto, R. Letor, and B. Ishan,  How short circuit capabilities
govern the desired characteristics of IGBT s, presented at the Power
(f) Conversion Conf., Apr. 1992.
[3] S. Clemente, A. Dubhashi, and B. Pelly,  IGBT characteristics and
applications, International Rectifier, El Segundo, CA, Application Note
AN-983.
At turn-on (see Figs. 8 and 9), as an IGBT is turned on, the
[4] R. Chokhawala, J. Catt, and L. Kiraly,  A discussion on IGBT short
switching causes voltage at module terminals
circuit Behavior and fault protection schemes, presented at the IEEE-
APEC Conf., Boston, MA, Feb. 23 27, 1992.
to drop from its initial value of to an amount equal to
[5] R. Chokhawala and G. Castino  IGBT fault current limiting circuit,
As explained in this paper, this causes the snubber
presented at the IEEE-IAS Annu. Meeting, Toronto, Ont., Canada, Oct.
3 8, 1993.
capacitors to discharge through  s.
[6] G. Castino, A. Dubhashi, S. Clemente, and B. Pelly,  Protecting
For simplicity, it is assumed that the turn-on is linear.
IGBT s against short circuit, International Rectifier, El Segundo, CA,
It is further assumed that the peak turn-on current is 25% above Application Note AN-984, 1989.
[7] T. Undeland et al.,  A snubber configuration for both power transistors
the load current at the switching instant. These assumptions
and GTO PWM inverters, in Conf. Rec. 15th Annu. IEEE-PESC Conf.,
will result in a conservative estimate of snubber losses. pp. 42 53.
1610 IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 33, NO. 6, NOVEMBER/DECEMBER 1997
Rahul S. Chokhawala (M 87) received the M.Tech Saed Sobhani received the B.S. degree in electrical
degree in power electronics from the Indian Institute engineering, with an emphasis in microelectronics,
of Technology, Madras, India, and the M.S. degree from the University of Colorado, Colorado Springs,
in control systems from the University of Iowa, in 1991.
Iowa City. In 1992, he joined International Rectifier Corpo-
He joined International Rectifier Corporation in ration, El Segundo, CA, where, as an Applications
1982 and worked in evaluations and applications Engineer, he performed extensive evaluations of
positions. He participated in the development of the company s IGBT power modules versus the
inverter SCR s, GTO s, HV snubber diodes, and competition to establish a competitive position in
IGBT modules. As a Senior Applications Engineer, the marketplace. He is currently a Senior Design
he researched applications-related issues for IGBT Engineer in the Research and Development Group,
products. He has published several papers in this area. He joined Motorola involved in development of a new generation of power devices, including
Power Products in March 1994 and was a Program Manager for IGBT temperature and current sensor designs that provide value-added solutions to
products. In this position, he worked to develop newer generations of motor the industry.
control IGBT s/FWD s and helped establish manufacturing capabilities for
discrete IGBT products. He is currently with ABB Semiconductors AG,
Lenzburg, Switzerland, where he is a Strategic Business Manager for power
systems products.