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VCO Designs for Wireless Handset and
CATV Set-Top Applications

APN1012

Introduction

Voltage Controlled Oscillators (VCOs) have come to the
forefront of RF designs together with the first PLL circuits.
In the era before the PLL, oscillators were mostly free
running and only in rare cases were varactors used for
modulation or temperature compensation. Nowadays, we
rarely see free running oscillators, instead they have
become varactor controlled oscillators. This is because
most RF applications require band coverage, which can
be realized through the PLL circuit requiring two sources
of RF power. The reference source frequency is often a
VCXO or TCXO, while the other frequency is controlled
by the PLL phase detector.

Usually, both VCXO/TCXO and the RF VCO are voltage
controlled oscillators. The difference between a reference
oscillator and a tuned VCO is that the former usually has
a very high-Q resonator, which allows for very stable

Application Note

oscillation, while the latter has a lower-Q resonator,
allowing a relatively high tuning range. In reference
oscillators, varactors are used for fine tuning or
temperature compensation (TCXO). In tunable oscillators,
varactors are used to change (tune) the frequency. In
some VCOs, varactors may be used also for modulation,
for example in a DECT system where modulation is used
to generate a constant-envelope GMSK signal.

Although it is a small part of the RF design, the VCO is
often a major headache for designers. The goal, in this
application note, is to show how Alpha’s products and
services may help you to overcome VCO concerns and
help to make your design among the best products
on the market.

2

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VCOs in Digital Wireless Phones

Consider the hypothetical wireless handset phone.
Today, the handset is a dual-band (cellular/PCS) and multi-
mode system employing many VCO functions. There are
many ways to realize these functions, making it virtually
impossible to specify the frequency and tuning range for
all designs. However, there are certain common features
that are outlined in Figure 1.

In a typical receiver, dual conversion superheterodyne
solutions are usually employed. They convert either
900 MHz (cellular) or 1.8 GHz (PCS) down to the SAW
frequency range, which may be between 90–400 MHz.

Further, this signal is either down-converted or
demodulated into a digital I/Q signal using a lower
frequency IF VCO. The transmitter path is either directly
modulated at 900 MHz or uses a dual conversion scheme
requiring at least two VCOs.

When dual-band requirements are needed, up to 8 or
more VCOs may be required to satisfy specific frequency
plans. This is often a technically and economically
restrictive solution. Many designers try to solve this over-
VCOed problem using both smart frequency planning and
multi-band VCOs, as shown in Figure 1.

VCO Designs for Wireless Handset and CATV Set-Top Applications

APN1012

RF VCO Ranging:

400–1900 MHz

T/R

Switch

I

IF VCO

RF VCO

BPF

BPF

BPF

BPF

LNA

RF

Detector

PA

BPF

PLL

Control

PLL

PLL

Control

PLL

π/2

Σ

BPF

PLL

Control

π/2

PLL

Control

PLL

I

Q

Q

RF VCO

IF VCO

Ranging:

100–400 MHz

PLL

Coupler

Figure 1. VCOs in a Digital Wireless Phone

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Fundamental Low Noise Colpitts VCO

The characteristic feature of the Colpitts VCO is that it uses
a capacitive divider for the feedback consisting of C

1

and

C

2

, and an inductive branch including a parallel resonator

and series capacitor C

3

. The parallel resonator includes

inductive element M

1

(that may be a discrete inductor for

lower frequencies or a length of micro-strip line for RF) and
a capacitive branch, consisting of a varactor and a series
capacitor(s). The entire inductive branch should have
inductive impedance at the frequency of oscillation,
otherwise there will be no oscillation. This means that the
resonant frequency should be higher than the oscillation
frequency.

Note that the resonator current circulates through the
varactor, series capacitor C

11

and inductor M

1

and is the

largest current in the tank circuit. Because of this, losses
introduced in this current path are the crucial ones with
respect to phase noise.

Without delving deeply into phase noise theory, we note
that phase noise is inversely proportional to the power
bypassed through the feedback loop, and the loaded Q
of the tank circuit. Thus, the more power lost on the way
to the transistor base, the higher the noise. It is clear that
varactor loss plays a crucial role in the phase noise
property of the VCO. If phase noise is an issue, the
varactor series resistance should be carefully considered.

There is an additional concern because phase noise is not
only a function of varactor loss. The varactor capacitance
voltage characteristic has a crucial impact on phase noise
as well. With a higher capacitance ratio, the varactor’s
coupling to the resonator is reduced resulting in lower
resonator current. Therefore, a hyperabrupt varactor
having higher series resistance is often a better choice
than a lower capacitance ratio abrupt varactor having lower
series resistance.

VCO Designs for Wireless Handset and CATV Set-Top Applications

APN1012

Q

1

2SC5007 (“34”)

Q

2

2SC5008 (“44”)

C

1

2 p

C

2

1.5 p

C

3

0.75 p

C

7

0.5 p (0402)

R

1

150

R

2

2.7 k

C

4

0.75 p

C

8

R

3

200

R

4

3.6 k

C

5

0.75 p

C

6

0.75 p

M

1

4 x 04 mm (Trim)

D

1

SMV1493

C

10

15 p

C

11

1.5 p (0603)

C

9

C

12

1 p

M

2

6.5 x 0.2 mm

M

3

4 x 0.35 mm

V

TUNE

M

4

2 x 0.2 mm

M

6

5 x 0.2 mm

M

5

2 x 0.2 mm

RF Out

V

CC

Low R

S

, Low Voltage

Hyperabrupt Varactor

Figure 2. Low Noise High Performance Colpitts VCO

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Differential VCO for Integration with
a RF IC

Designs based on RF IC solutions, with built-in VCOs,
often employ a differential VCO configuration. One
possible differential VCO configuration is shown in Figure
3. In this case, the tank circuit is formed by C

3

, C

4

and a

resonator C

8

, C

9

, D

1

, M

1

. Here again, the resonator

current plays a decisive role in phase noise definition.
Thus, phase noise is strongly dependent on resonator
loss. Capacitors C

3

and C

4

help establish the correct

phase shift value in the feedback loop moving oscillations

closer to the resonant frequency. This is the principal
difference between a Colpitts and a differential VCO. In
the Colpitts case, the resonant frequency is always higher
than the oscillation frequency; in the differential VCO the
resonant and oscillation frequencies may coincide. Thus
the loaded Q of the circuit becomes significantly higher,
and the feedback loop losses are increased due to the
higher resonant currents. When this happens, the
differential VCO is more vulnerable to resonator loss than
the Colpitts VCO and usually shows 5–10 dB higher noise
if compared to an equivalent Colpitts case.

VCO Designs for Wireless Handset and CATV Set-Top Applications

APN1012

V

1

V

2

V

3

R

1

47

R

2

3 k

R

3

9.1 k

R

4

3 k

R

5

3 k

R

6

3 k

R

7

3 k

R

8

20

R

9

3.9 k

L

1

33 n

C

1

100 p

C

2

2 p

C

3

4 p

C

4

1.5 p

D

1

SMV1493-079

C

5

1.5 p

C

6

100 p

C

7

100 p

V

CC

+3 V

V

VAR

RF Out

C

8

2 p

C

9

6 p

M

1

4 x 0.5 mm

M

2

6.5 x 0.2 mm

R

10

51

Low R

S

, Low Voltage

Hyperabrupt Varactor

RF IC

Figure 3. Differential VCO for the Integration with the RF IC

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Dual-Band Switchable VCO Schematic

One way to improve design economics in the multi-VCO
requirement is to employ band switching in the VCO. If the
frequency switching required isn’t very large (say within
20%) it may usually be realized within the same tank
circuit, by switching “on” or “off” an additional capacitor or
inductor. However, if the required switching is more than
30%, it becomes very difficult to satisfy both wideband and
low noise requirements in a single design. One possible
solution is to use two separate tank resonator circuits
switched with two PIN diodes. In this case, the feedback
needs to be optimized to fit both band requirements at the
same time. Thus, a trick is used — connecting a capacitor

C

11

in parallel with C

6

when a lower-band resonator is

selected. This provides significant improvement in phase
noise since C

6

may then be optimized for the best

performance at high band, and C

11

at the lower band.

Another important feature of this switching scheme is that
the PIN diodes are not in the resonator current path.
Because of this, phase noise is not sensitive to the PIN
diode resistance. This is fortunate, since it means less
forward current is needed. In addition, any noise on the
PIN diode bias current (common for the noisy digital
environment of today’s phones) would not cause
significant modulation noise.

VCO Designs for Wireless Handset and CATV Set-Top Applications

APN1012

L

1

12 nH

D

1

C

1

8 pF

C

2

470 pF

C

3

100 pF

C

6

20 pF

C

7

15 pF

R

1

300

R

2

R

3

P

1

L

2

56 nH

D

2

SMV1139-011

C

8

10 pF

R

4

1.5 k

P

2

SMP1320-011

1

J

1

V

TUNE

1

J

2

V

SW_High

1

J

3

V

SW_Low

C

4

100 pF

C

5

100 pF

Q

1

NE68519

C

9

100 pF

C

10

20 pF

R

5

100

R

6

3 k

R

7

6.8 k

1

J

4

V

CC

+3 V

1

J

5

RF

1.5 k

1.5 k

C

11

30 pF

C

CC

100 pF

SMV1408-011

SMP1320-011

Low Current Switching

PIN Diodes

Low R

S

, Low Voltage

Hyperabrupt Varactor

Figure 4. Dual-Band Switchable VCO Schematic

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Dual-Band Switchable RF VCO

As mentioned before, relatively small (less than 20%)
frequency switching may be achieved inside the same tank
circuit by connecting or disconnecting capacitors (and
sometimes inductors). The PIN diode D

2

performs a tricky

task ¾ it adds more capacitance in parallel with the
existing parallel capacitance of the resonator, and also
adds more capacitance in parallel with the existing series

capacitor. This technique is used to overcome the problem
of increased resonator Q, when connecting additional
parallel capacitance, by decreasing it with higher series
capacitance. It allows D

2

to keep phase noise near its

optimum at both bands. Another PIN diode in the output
matching circuit tunes the buffer to a frequency doubler
mode when working in PCS band.

VCO Designs for Wireless Handset and CATV Set-Top Applications

APN1012

V

CC

(3 V)

RF Out

D

3

D

2

D

1

V

VAR

V

CTL1

V

CTL2

D

1

: SMV1493-079

D

2

, D

3

: SMP1322-079

PINs are Using in Both Resonator Tank

and Output Matching Circuits

Figure 5. Dual-Band Switchable RF VCO

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VCOs in a Set-Top Cable Down-Converter

The typical set-top down-converter is a dual-conversion
receiver employing up-conversion and down-conversion
techniques to overcome image problems in a wideband
environment of 50–1000 MHz. In the dual up-/down-
conversion scheme, the problem of image channel and
input filtering virtually does not exist because there is no
signal at the image channel. The image channel is always
higher than the highest frequency of the cable signal.

Two RF VCOs are required for dual down conversion. The
first is a wideband VCO tuned from 1100–2000 MHz with
a control voltage from 1–20 V. The other is a narrow band
VCO, which may use a CDR, coaxial dielectric resonator,
at 1144 MHz. In a digital system the second IF signal
may be further demodulated, requiring an additional
44 MHz VCO.

The specific action of the wideband VCO is its wideband
tuning requirement. Let us consider some possible
solutions for the wideband VCO.

VCO Designs for Wireless Handset and CATV Set-Top Applications

APN1012

BPF

PLL

PLL

LPF

HPF

LPF

Upstream

filter

BPF

Upstream

PLL

Control

PLL

Control

AGC

1st Mixer

75

Ω

54–860 MHz

40 dB

45.75/44 MHz

1100 MHz

1154–1960 MHz

1144/1145 MHz

2nd Mixer

Low-Distortion PIN Diode

Attenuator

Wideband VCO Tuning in

1100–2000 MHz Range

Narrow-Band VCO

@ 1145 MHz

dB

Figure 6. VCOs in a Set-Top Cable Down-Converter

8

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Wideband Colpitts VCO Schematic

The unique action of the wideband Colpitts VCO is in its
tank circuit design, which uses an inductor with a varactor
connected in series and no parallel capacitor in contrast
to the low noise Colpitts VCO described in Figure 7. The
feedback capacitors are optimized to the best power

flatness over the entire frequency band. Back-to-back
varactors are often used to minimize parasitic mounting
capacitance (between mounting pads and adjoining
components). This circuit is usually designed to minimize
any parasitic parallel capacitance that may be caused by
component pads or transmission lines close to the
inductive path.

VCO Designs for Wireless Handset and CATV Set-Top Applications

APN1012

V

CC

= 5 V

I

CC

= 9 mA

NE68519

SMV1265-011

3.3 k

9.1 k

200

3 k

3 k

320 x 30 mils

1 p

560 p

1.62 p

300 p

2 p

5.6 nH

SMV1265-011

RF Output

V

TUNE

High C (V) Ratio

Hyperabrupt Varactors

Useful Tuning

Range:

980–2120 MHz

0.9

1.0

1.1

1.2

1.3

1.4

1.5

1.6

1.7

1.8

1.9

2.0

2.1

2.2

0

5

10

15

20

25

30

Varactor Voltage (V)

Frequency (GHz)

P

OUT

(dBm)

0

1

2

3

4

5

6

7

Fexp

Fmodel

P

OUT

_exp

P

OUT

_model

A carefully designed layout with minimum parasitic
capacitances may show large frequency coverage, for
example 980–2120 MHz as the performance indicates.

The varactor selection is a crucial part of the design.
Alpha’s new SMV1265-011 varactor is specifically
designed to fit this wideband application.

Figure 7. Wideband Colpitts VCO Schematic

Figure 8. Wideband Colpitts VCO Performance

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Wideband Balanced VCO Schematic

An even wider tuning range may be achieved with a
balanced VCO configuration. The reason for its wider
tuning performance is that the phase response of this
VCO’s active element is flatter over the range of tuning

compared to a Colpitts VCO. This allows the tank circuit
more control over the oscillation frequency.

The best results are achieved with back-to-back
connected SMV1265 varactors, where there is
820–2120 MHz coverage.

VCO Designs for Wireless Handset and CATV Set-Top Applications

APN1012

V

1

NE68119

V

2

NE68119

T1

16 x 0.4 mm

T2

15 x 0.7 mm

C

1

10

C

2

10

L

1

33 nH

L

2

33 nH

R

1

33

R

2

33

R

3

120

R

4

120

R

5

820

R

7

51

R

8

1 k

R

6

820

R

10

2.4 k

D

1

R

9

1 k

R

11

1000

C

3

100

V

VAR1

V

CC1

5–8 V

C

4

100

C

5

100

R

12

5

0

A

D

2

SMV1265

SMV1265

T3

3 X 0.7 mm

C

6

1000

High C (V) Ratio

Hyperabrupt Varactors

0

5

10

15

20

25

30

Varactor Voltage (V)

Varactor Voltage (V)

0.8

1.0

1.2

1.4

1.6

1.8

2.0

2.2

Frequency (GHz)

Frequency Tuning

0

5

10

15

20

25

30

-8

-6

-4

-2

0

2

4

6

8

Power (dBm)

Power Response

Useful Tuning

Range:

820–2120 MHz

Measured @ 7 V

Simulated @ 7 V

Measured @ 5 V

Measured

Simulations

Figure 9. Wideband Balanced VCO Schematic

Figure 10. Wideband Balanced VCO Performance

10

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Varactor Fundamentals

Let us consider some fundamental properties of
varactors. A varactor is a specially designed P-N junction
diode, whose capacitance changes significantly in
reverse bias mode. There are three important parameters
characterizing varactors. The first is the capacitance ratio
at two reverse voltages; this value characterizes the tuning
ability of the varactor capacitance and is one of the most
important parameters. The second is the value of
capacitance at a given voltage. The third is the series
resistance of the varactor.

The structure of the basic varactor, called an abrupt
junction varactor, is shown in Figure 11. Generally, it is built
as a P++ - N - N+ structure, using epitaxial N-growth on
the N+ substrate with a constant doping level in the N-
region. The lower doped N-region is the active area where
the electron concentration changes depending on the
reverse voltage applied between the anode and cathode
of the varactor. There are certain limitations on the level
of doping in the N-region, which is usually defined by the
required capacitance ratio of the varactor. Because of this,

the conductance of the N-area is a major contributor to
the varactor’s series resistance. Note that as the reverse
voltage is increased, the series resistance (due to the
N-area) will decrease along with the capacitance.

The hyperabrupt junction varactor has a more
complicated doping profile. Because of much higher
doping on the P++ border, the electron concentration
changes much more abruptly compared to an abrupt
junction. As a result, the capacitance of the hyperabrupt
diode at zero bias is much higher than for the abrupt diode.
Therefore, the capacitance change vs. reverse bias
becomes significantly higher for hyperabrupt diodes. The
trade-off for this better capacitance ratio is increased
series resistance. The reason is that the doping level of
the N-area has been reduced to keep average doping level
over the N-region the same as the abrupt diode level.
There are many ways to bring the series resistance in the
hyperabrupt diode to as low a level as possible. Modern
state-of-the-art hyperabrupt diodes for low noise VCOs
have series resistance almost as low as discrete
ceramic capacitors.

VCO Designs for Wireless Handset and CATV Set-Top Applications

APN1012

P++

Doping

Level

N

Depth from Anode

V

0

V

0

V

1

V

1

V

2

V

2

V

SAT

V

SAT

Electron Concentration

P++

Doping

Level

N+

N

Depth from Anode

Electron Concentration

Abrupt Junction

Hyperabrupt Junction

N+

Figure 11. Varactor Fundamentals

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Varactor Packaging

Most high-volume discrete applications require varactors
in low cost, small surface mount plastic packages. Alpha
Industries provides a large variety of both plastic and

ceramic packages. The recent most advanced miniature
plastic package, SC-79, shown in Figure 12, is as small
as 0402 discrete components.

VCO Designs for Wireless Handset and CATV Set-Top Applications

APN1012

24 mil (0.62 mm)

62 mil (1.58 mm)

SC-79

Figure 12. Varactor Packaging

The coupling coefficient may be derived from the known
(or typical) values of the tuning frequency and varactor
capacitance variation. Note that the total temperature drift
in this case is about 0.5%, as compared to 1% maximum

drift caused by temperature compensated discrete
ceramic capacitors. Even those numbers are extremely
small when compared to the temperature drifts caused by
a VCO transistor.

For the Typical Wireless Case:

f = 1.6 ± -0.04 GHz

Using SMV1235-011 Varactor:

The Total Temperature Drifts

Due to Varactor in the

-40 to +85˚C Becomes:

L

RES

C

RES

C

DIV1

C

DIV2

C

BE

C

CE

C

CB

C

SER2

C

VAR

0.24

8

1.6 GHz

0.08 GHz

2

2

≈

=

x

x

∆C

VAR

=

3.4 pF

pF

C

f

∆f

K

Compare to the Discrete Capacitor!

f

∆f

T

%

54

.

0

≈

12

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VCO Designs for Wireless Handset and CATV Set-Top Applications

APN1012

Relative Capacitance Change vs.
Temperature

Figure 13 shows typical relative capacitance variations vs.
temperature for different reverse voltages. It indicates a
total capacitance change of 5–6% in the range of -40°C
to +80°C. In comparison, a temperature compensated

ceramic capacitor residual variation bar is shown for a
typical ±100 ppm device. This has a possible total
capacitance change of over 2%. When comparing the
overall effect of temperature on varactors and ceramic
capacitors, the coupling of the devices to the resonator
circuit should be considered.

-4

-3

-2

-1

0

1

2

3

4

5

-40

-20

0

20

40

60

80

Temperature (˚C)

Percentage of Variation (%)

Consider

Varactor

Coupling!!

V

VAR

= -4 V

V

VAR

= -1 V

V

VAR

= -2.5 V

Deviation Range for a Typical

Temperature Compensated

Discrete Ceramic

Multilayer Capacitor

Figure 13. Relative Capacitance Change vs. Temperature for Hyperabrupt Varactors

Figure 14. Varactor Temperature Effect on the Oscillation Frequency

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VCO Designs for Wireless Handset and CATV Set-Top Applications

APN1012

M

+ C

P

V

J

V

VAR

C

GO

C

V

=

1 +

SPICE Model for SMV1142-011









Varactor SPICE Model

To model a varactor in most commercial simulators, we
recommend the available PN-junction diode SPICE model.
We specify the barrier junction capacitance parameters
C

GO

, V

J

, and M instead of the default parameters. In

addition, we add a value of C

P

, in parallel with the junction

capacitor, which is not the package capacitance. For ideal
abrupt junction varactors, the parameters are constant and
may be defined from physical theory. However, for actual
abrupt or hyperabrupt varactors, these values are not
constant. In these cases, we use the same equation, to
fit its parameters, for the best compliance with measured
capacitance vs. voltage response.

Because of formalization, parameters describing the
junction capacitance of hyperabrupt varactors may be
significantly different from the default values used in the
SPICE simulators for the ideal silicon PN-junction.
For example, typical hyperabrupt varactor SMV1235 was
fitted with M = 4 as opposed to 0.5, which follows silicon

PN diode theory. Note that some SPICE simulators offer
fixed default values of M = 0.5 which can’t be changed.
In this case, a diode model may not be used, however,
direct nonlinear capacitance may be used as defined in
the given formula.

0

2

4

6

8

10

12

Varactor Voltage

0

5

10

15

20

Capacitance (pF)

SMV1235
Approximation

SMV1235

16.13/(1-Vv/8)^4 + 2

8

1+

16.13

4

+ 2

=

V

VAR

C

V













Figure 15. Typical Varactor SPICE Model

Figure 16. C (V) Curve Fitting for Typical Hyperabrupt Varactor

14

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Super-hyperabrupt Varactor Modeling

To overcome limitations of the “standard” PN-junction
SPICE model for hyperabrupt and super-hyperabrupt
devices, such as the SMV1265, an interleaving technique
is used. In this technique, the entire capacitance reverse
voltage range is broken into several subranges. These

subranges are small enough for the formula to provide
good approximation not only within a given subrange, but
also for certain extensions beyond it. The extension margin
is defined from previously estimated RF signal amplitude.
Such interleaving ensures that the formula would work well
not only in terms of DC bias, but for large-signal RF
analysis as well.

VCO Designs for Wireless Handset and CATV Set-Top Applications

APN1012

M =

pwl (V

VAR

0 2 2.5 2 2.500009 25 6.5 25 6.50009 7.3 11 7.3 11.0009 1.8 40 1.8)

V

J

=

pwl (V

VAR

0 4 2.5 4 2.500009 68 6.5 68 6.50009 14 11 14 11.0009 1.85 40 1.85)

C

P

=

pwl (V

VAR

0 0 2.5 0 2.500009 0 6.5 0 6.50009 0.9 11 0.9 11.0009 0.56 40 0.56)

C

JO

=

pwl (V

VAR

0 22.5 2.5 22.5 2.500009 21 6.5 21 6.50009 20 11 20 11.0009 20 40 20)*10

-12

R

S

=

pwl (V

VAR

0 2.4 3 2.4 4 2.3 5 2.2 6 2 7 1.85 8 1.76 9 1.7 10 1.65 11 1.61 12 1.5 40 1.5)

0

5

10

15

20

25

30

Varactor Voltage

0.1

1.0

10

100

Capacitance (pF)

0

0.2

0.4

0.6

0.8

1.0

Approximation

Measured

SMV1265

Interleaving of Splines

Figure 17. Piece-Wise Curve Fitting for High C (V) Ratio Varactors

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VCO Modeling Concept

For the purpose of modeling and analysis, a VCO design
may be simulated as an amplifier with parallel feedback.
This analysis involves measuring loop gain using a specific
idealized directional coupler called “OSCTEST” in

Libra IV. (For Harmonica users there is an application note
showing how to implement OSCTEST function using
S-parameters file. Refer to your Harmonica vendor for
more information).

VCO Designs for Wireless Handset and CATV Set-Top Applications

APN1012

RL

L

C

C1

C2

I1

I2

V1

V2

Simplified Colpitts VCO

Feedback Model of Colpitts VCO

Amplifier

Tank Circuit

Loop Gain

Observation

Plain

Figure 18. VCO Modeling Concept

16

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VCO Designs for Wireless Handset and CATV Set-Top Applications

APN1012

The major goal of the large-signal open loop VCO analysis
is to observe the magnitude (defined in dB) and the phase
of the open loop voltage gain Ku, to identify particular
features of the designed VCO.

First, we need to establish the optimum conditions for the
oscillations in a given tuning range. Second, we need to
find out whether there are possibilities for parasitic

oscillations both in the lower and higher frequency ranges.
If there are parasitic oscillations, some preventive
measures should be taken. Third, we need to find ways
to make both Q

L

and the loop power (P

IN

) as high as

possible to facilitate phase noise performance. Finally,
other features of the VCO need to be addressed, among
them load pulling and V

CC

pushing.

1.0

1.5

2.0

2.5

3.0

3.5

Frequency (GHz)

-4

-3

-2

-1

0

1

2

3

4

Mag (Ku) dB

-200

-100

0

100

200

Oscillation

Point

Arg (Ku)

Mag (Ku)

0.5

1.0

1.5

2.0

2.5

Frequency

-250

-200

-150

-100

-50

50

0

100

V

VAR

= 0 V

Transitor X at P

IN

= 10 dBm

Resonator at Different

Varactor Voltages

Parallel Resonance:

When it nears the oscillation

point, tank circuit losses

increase; noise increase and

power decrease follows

Oscillation Happens at the 0 dB Gain

and 0 Loop Phase Shift

V

VAR

= 30 V

Figure 19. Typical Loop Gain Results for the Colpitts VCO

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VCO Designs for Wireless Handset and CATV Set-Top Applications

APN1012

Wideband Colpitts VCO Model

The OSCTEST component interrupts the oscillator
feedback, allowing the designer to analyze the VCO as an
ordinary two-port circuit (amplifier). To observe the loop
response, we define the open loop voltage gain Ku. For

more details, please refer to the VCO application notes
listed in the References section. The varactor model is
defined as a PN-junction diode SPICE model for large
signal harmonic balance analysis. The transistor is
described by the Gumel Poon SPICE model with
parameters provided by the vendor.

Transistor
Subcircuit

Seamless

VCO Loop

Opener

Varactor

Model

Figure 20. Wideband Colpitts VCO Model

18

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VCO Designs for Wireless Handset and CATV Set-Top Applications

APN1012

Differential VCO Fundamentals

The differential VCO utilizes paired transistors in
common-emitter and common-base configurations. The
phase balance condition for sustaining oscillations
requires significantly lower phase shift in comparison to

a Colpitts design (ideally 0 degrees vs. 180 degrees). This
makes it possible to use a resonator tuned to the exact
resonant frequency. However, the feedback losses may be
higher because the higher resonating currents will cause
increased ohmic losses.

Resonator Works at it’s

Parallel Resonance, Giving

Best Phase Slope Performance

Two Transistors in the

Loop Give Advantage

of Higher Loop Gain

Common-collector -

Common-base Phase

Shift Ideally Is 0

Figure 21. Concept of Differential VCO

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VCO Designs for Wireless Handset and CATV Set-Top Applications

APN1012

Balanced VCO Fundamentals

Fundamental properties of the balanced VCO are more
clearly understood using the simplified circuit diagram
shown in Figure 22. The transistors are in common
collector configuration. This is characterized by high input
impedance, looking from the transmission line and
referenced as L

B

. Capacitor C

BP

simulates the

transmission lines and the grounding effect of the
mounting pads. Coupling current ICPL circulates between
the transistor bases to drive them with a 180° phase shift.

The emitter current I

FB

forms the feedback loop, carrying

an amplified energy surplus that is needed to sustain
resonant current I

RES

and coupling current ICPL through

the emitter-base path. Unlike a Colpitts VCO, this circuit
does not require frequency dependent feedback to match
the internal transistor high frequency phase shifts. When
properly compensated for wideband performance with
inter-base inductor, L

B

, this circuit will be more broadband

than a Colpitts VCO.

L

PAR

L

SER

D

VAR

Re2

Le2

C

BP

C

BP

Re1

Le1

L

B

Rcol

C

BP

C

BP

Rcol

I

CPL

I

FB

I

RES

Low-pass Matching

Serves to Improve

High-frequency

Performance

Collector Currents

are Shifted 180˚

In Phase. That’s

Why We Call It

“Balanced”

Figure 22. Balanced VCO Fundamentals

References

“Varactor SPICE Models for RF VCO Applications.”
Applications Note APN1004, Alpha Industries, Inc., 1998.

“A Colpitts VCO for Wideband (0.95 GHz–2.15 GHz) Set-
Top TV Tuner Applications.” Applications Note APN1006,
Alpha Industries, Inc., 1998.

“A Balanced Wideband VCO for Set-Top TV Tuner
Applications.” Applications Note APN1005, Alpha
Industries, Inc., 1998.

“Switchable Dual-Band 170/420 MHz VCO For Hand-Set
Cellular Applications.” Applications Note APN1007,
Alpha Industries, Inc., 1998.

“A Wideband General Purpose PIN Attenuator.”
Applications Note APN1003, Alpha Industries, Inc., 1999.

“Wideband VCO for Set-Top Applications.” Microwave
Journal, April 1999.

“Circuit Models for Plastic Packaged Microwave Diodes.”
Applications Note APN1001, Alpha Industries, Inc.

“Design with PIN Diodes.” Applications Note APN1002,
Alpha Industries, Inc.

For the availability of the above materials, visit Alpha
Industries Web site at: www.alphaind.com.